Elan Microelectronics Corporation v. Apple, Inc.

Filing 237

Appendix A & B [AND EXHIBITS 1-8] TO THE DECLARATION OF RAVIN BALAKRISHNAN IN SUPPORT OF APPLE INC'S OPPOSITION TO ELAN MICROELECTRONICS CORP.'S MOTION FOR PARTIAL SUMMARY JUDGMENT OF INFRINGEMENT OF U.S. PATENT NO. 5,825,352 filed byApple, Inc.. (Attachments: # 1 Appendix B, # 2 Exhibit 1, # 3 Exhibit 2, # 4 Exhibit 3, # 5 Exhibit 4, # 6 Exhibit 5, # 7 Exhibit 6, # 8 Exhibit 7, # 9 Exhibit 8)(Greenblatt, Nathan) (Filed on 6/2/2011)

Exhibit 6

111111111111111111111111111111111111111111111111111111111111111111111I11111
US0054950nA

United States Patent

[11]

Miller et al.
[54]

OBJECT POSITION AND PROXIMITY
DETECTOR

[75]

Inventors: Robert J. Miller, Fremont; Stephen
Bisset, Palo Alto; Timothy P. Allen,
Los Gatos; Gunter Steinbach, Palo
Alto, all of Calif.

Date of Patent:

5,495,077

Patent Number:

[45]

[19]

* Feb. 27, 1996

References Cited

[56]

U.S. PATENT DOCUMENTS

[73]

Notice:

The portion of the term of this patent
subsequent to Dec. 20, 2011, has been
disclaimed.

[21]

Appl. No.: 252,969

[22]

Filed:

Jun. 2, 1994
Related U.S. Application Data

[63]

Continuation of Ser. No. 115,743, Aug. 31, 1993, Pat. No.
5,374,787, which is a continuation-in-part of Ser. No. 895,
934, Jun. 8, 1992.

[51]
[52]
[58]

Int. Cl. 6
U.S. CI
Field of Search

32-1

G08C 21/00; H03M 11100
178/18; 345/173; 341/33

ABSTRACT

[57]

A proximity sensor system includes a sensor matrix array
having a characteristic capacitance on horizontal and vertical conductors connected to sensor pads. The capacitance
changes as a function of the proximity of an object or objects
to the sensor matrix. The change in capacitance of each node
in both the X and Y directions of the matrix due to the
approach of an object is converted to a set of voltages in the
X and Y directions. These voltages are processed by analog
circuitry to develop electrical signals representative of the
centroid of the profile of the object, i.e, its position in the X
and Y dimensions. The profile of position may also be
integrated to provide Z-axis (pressure) information.

178/18; 345/173-178;
341/33,34
34-1

38

\

178/18

Primary Examiner-Stephen Chin
Assistant Examiner-Kevin Kim
Attorney, Agent, or Firm-D' Alessandro & Ritchie

Assignee: Synaptics, Inc., San Jose, Calif.

[*]

5,374,787 12/1994 Miller et aI

11 Claims, 12 Drawing Sheets
40

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Feb. 27, 1996

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Sheet 1 of 12

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Feb. 27, 1996

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Feb. 27, 1996

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5,495,077
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2

phenomena and may be sensltive to the user's rate of
movement. In addition, strain gauge or pressure plate
approaches are a somewhat expensive because special sensors are required.
RELATED APPLICATIONS
Optical approaches are also possible but are somewhat
5
limited for several reasons. All would require light generaThis application is a continuation of application Ser. No.
tion which will require external components and increase
081115,743, filed Aug. 31, 1993, now U.S. Pat. No. 5,374,
cost and power drain. For example, a "finger-breaking"
787, which is a continuation-in-part of co-pending applicainfra-red matrix position detector consumes high power and
tion Ser. No. 7/895,934, filed Jun. 8, 1992 and assigned to
10 suffers from relatively poor resolution.
the same assignee as the present invention.
There have been numerous attempts to provide a device
for sensing the position of thumb or other finger for use as
BACKGROUND OF THE INVENTION
a pointing device to replace a mouse or trackball. Desirable
attributes of such a device are low power, low profile, high
1. Field of the Invention
15 resolution, low cost, fast response, and ability to operate
The present invention relates to object position sensing
reliably when the finger carries electrical noise, or when the
transducers and systems. More particularly, the present
touch surface is contaminated with dirt or moisture.
invention relates to object position recognition useful in
Because of the drawbacks of resistive devices, many
applications such as cursor movement for computing
attempts have been made to provide pointing capability
devices and other applications.
20 based on capacitively sensing the position of the finger. U.S.
2. The Prior Art
Pat. No. 3,921,166 to Volpe teaches a capacitive matrix in
Numerous devices are available or have been proposed
which the finger changes the transcapacitance between row
for use as object position detectors for use in computer
and column electrodes. U.S. Pat. No. 4,103,252 to Bobick
systems and other applications. The most familiar of such
employs four oscillating signals to interpolate x and y
devices is the computer "mouse". While extremely popular 25 positions between four capacitive electrodes. U. S. Pat. No.
as a position indicating device, a mouse has mechanical
4,455,452 to Schuyler teaches a capacitive tablet wherein
parts and requires a surface upon which to roll its position
the finger attenuates the capacitive coupling between elecball. Furthermore, a mouse usually needs to be moved over
trodes.
long distances for reasonable resolution. Finally, a mouse
30
U.S. Pat. No. 4,550,221 to Mabusth teaches a capacitive
requires the user to lift a hand from the keyboard to make the
tablet wherein the effective capacitance to "virtual ground"
cursor movement, thereby upsetting the prime purpose,
is measured by an oscillating signal. Each row or column is
which is usually typing on the computer.
polled sequentially, and a rudimentary form of interpolation
Trackball devices are similar to mouse devices. A major
is applied to resolve the position between two rows or
difference, however is that, unlike a mouse device, a track- 35 columns. An attempt is made to address the problem of
ball device does not require a surface across which it must
electrical interference by averaging over many cycles of the
be rolled. Trackball devices are still expensive, have moving
oscillating waveform. The problem of contamination is
parts, and require a relatively heavy touch as do the mouse
addressed by sensing when no finger was present, and
devices. They are also large in size and doe not fit well in a
applying a periodic calibration during such no-finger-present
volume-sensitive application like a laptop computer.
40 periods. U.S. Pat. No. 4,639,720 to Rympalski teaches a
There are several available touch-sense technologies
tablet for sensing the position of a stylus. The stylus alters
which may be employed for use as a position indicator.
the transcapacitance coupling between row and column
Resistive-membrane position sensors are known and used in
electrodes, which are scanned sequentially. U.S. Pat. No.
several applications. However, they generally suffer from
4,736,191 to Matzke teaches a radial electrode arrangement
poor resolution, the sensor surface is exposed to the user and 45 under the space bar of a keyboard, to be activated by
is thus subject to wear. In addition, resistive-membrane
touching with a thumb. This patent teaches the use of total
touch sensors are relatively expensive. A one-surface
touch capacitance, as an indication of the touch pressure, to
approach requires a user to be grounded to the sensor for
control the velocity of cursor motion. Pulsed sequential
reliable operation. This cannot be guaranteed in portable
polling is employed to address the effects of electrical
computers. An example of a one-surface approach is the 50 interference.
UnMouse product by MicroTouch, of Wilmington, Ma. A
U.S. Pat. Nos. 4,686,332 and 5,149,919, to Greanias,
two-surface approach has poorer resolution and potentially
teaches a stylus and finger detection system meant to be
will wear out very quickly in time.
mounted on a CRT. As a finger detection system, it's XJY
Resistive tablets are taught by U.S. Pat. No. 4,680,430 to
sensor matrix is used to locate the two matrix wires carrying
Yoshikawa, U.S. Pat. No. 3,497,617 to Ellis and many 55 the maximum signal. With a coding scheme these two wires
others. The drawback of all such approaches is the high
uniquely determine the location of the finger position to the
power consumption and the high cost of the resistive memresolution of the wire stepping. For stylus detection, Grebrane employed.
anias first coarsely locates it, then develops a virtual dipole
by driving all lines on one side of the object in one direction
Surface Acoustic Wave (SAW) devices have potential use
as position indicators. However, this sensor technology is 60 and all lines on the opposite side in the opposite direction.
This is done three times with different dipole phases and
expensive and is not sensitive to light touch. In addition,
signal polarities. Assuming a predetermined matrix response
SAW devices are sensitive to residue buildup on the touch
to the object, the three measurements present a set of
surfaces and generally have poor resolution.
simultaneous equations that can be solved for position.
Strain gauge or pressure plate approaches are an interestU.S. Pat. No. 4,733,222 to Evans is the first to teach a
ing position sensing technology, but suffer from several 65
capacitance touch measurement system that interpolates to a
drawbacks. This approach may employ piezo-electric transhigh degree. Evans teaches a three terminal measurement
ducers. One drawback is that the piezo phenomena is an AC

OBJECT POSITION AND PROXIMITY
DETECTOR

5,495,077

3

4

system that uses a drive, sense and electrode signal set (3
signals) in its matrix, and bases the measurement on the
attenuation effect of a finger on the electrode node signal
(uses a capacitive divider phenomena). Evans sequentially
scans thru each drive set to measure the capacitance. From
the three largest responses an interpolation routine is applied
to determine finger position. Evans also teaches a zeroing
technique that allows "no-finger" levels to be cancelled out
as part of the measurement.
U.S. Pat. No. 5,016,008 to Gruaz describes a touch
sensitive pad that also uses interpolation. Gruaz uses a drive
and sense signal set (2 signals) in the touch matrix and like
Evans relies on the attenuation effect of a finger to modulate
the drive signal. The touch matrix is sequentially scanned to
read each matrix lines response. An interpolation program
then selects the two largest adjacent signals in both dimensions to determine the finger location, and ratiometrically
determines the effective position from those 4 numbers.
Gerpheide, PCT application US90/04584, publication No.
W091/03039, applies to a touch pad system a variation of
the virtual dipole approach of Greanias. Gerpheide teaches
the application of an oscillating potential of a given frequency and phase to all electrodes on one side of the virtual
dipole,and an oscillating potential of the same frequency and
opposite phase to those on the other side. Electronic circuits
develop a "balance signal" which is zero when no finger is
present, and which has one polarity if a finger is on one side
of the center of the virtual dipole, and the opposite polarity
if the finger is on the opposite side. To acquire the position
of the finger initially, the virtual dipole is scanned sequentially across the tablet. Once the finger is located, it is
"tracked" by moving the virtual dipole toward the finger
once the finger has moved more than one row or column.
Because the virtual dipole method operates by generating
a balance signal that is zero when the capacitance does not
vary with distance, it only senses the perimeter of the finger
contact area, rather than the entire contact area. Because the
method relies on synchronous detection of the exciting
signal, it must average for long periods to reject electrical
interference, and hence it is slow. The averaging time
required by this method, together with the necessity to
search sequentially for a new finger contact once a previous
contact is lost, makes this method, like those before it, fall
short of the requirements for a fast pointing device that is not
affected by electrical interference.
It should also be noted that all previous touch pad
inventions that used interpolation placed rigorous design
requirements on their sensing pad. Greanias and Evans use
a complicated and expensive drive, sense and electrode line
scheme to develop their signal. Gruaz and Gerpheide use a
two signal drive and sense set. In the present invention the
driving and sensing is done on the same line. This allows the
row and column sections to be symmetric and equivalent.
This in tum allows independent calibration of all signal
paths, which makes board layout simpler and less constraining, and allows for more unique sensor topologies.
The shortcomings of the inventions and techniques
described in the prior art can also be traced to the use of only
one set of driving and sensing electronics, which was
multiplexed sequentially over the electrodes in the tablet.
This arrangement was cost effective in the days of discrete
components, and avoided offset and scale differences among
circuits.
The sequential scanning approach of previous systems
also made them more susceptible to noise. Noise levels
could change between successive measurements, thus

changing the measured signal and the assumptions used in
interpolation routines.
Finally, all previous approaches assumed a particular
signal response for finger position versus matrix position.
Because the transfer curve is very sensitive to many parameters and is not a smooth linear curve as Greanias and
Gerpheide assume, such approaches are limited in the
amount of interpolation they can perform.
It is thus an object of the present invention to provide a
two-dimensional capacitive sensing system equipped with a
separate set of drive/sense electronics for each row and for
each column of a capacitive tablet, wherein all row electrodes are sensed simultaneously, and all column electrodes
are sensed simultaneously.
It is a further object of the present invention to provide an
electronic system that is sensitive to the entire area of
contact of a finger with a capacitive tablet, and to provide as
output the coordinates of some measure of the center of this
contact area while remaining insensitive to the characteristic
profile of the object being detected.
It is a further object of the present invention to provide an
electronic system that provides as output some measure of
area of contact of a finger with a capacitive tablet.

5

10

15

20

25

BRIEF DESCRIPTION OF THE INVENTION

30

35

40

45

50

55

60

65

With the advent of very high levels of integration, it has
become possible to integrate many channels of driving/
sensing electronics into one integrated circuit, along with the
control logic for operating them, and the interface electronics to allow the pointing device to communicate directly
with a host microprocessor. The present invention uses
adaptive analog techniques to overcome offset and scale
differences between channels, and can thus sense either
transcapacitance or self-capacitance of all tablet rows or
columns in parallel. This parallel-sensing capability, made
possible by providing one set of electronics per row or
column, allows the sensing cycle to be extremely short, thus
allowing fast response while still maintaining immunity to
very high levels of electrical interference.
The present invention comprises a position-sensing technology particularly useful for applications where finger
position information is needed, such as in computer
"mouse" or trackball environments. However the positionsensing technology of the present invention has much more
general application than a computer mouse, because its
sensor can detect and report if one or more points are being
touched. In addition, the detector can sense the pressure of
the touch.
According to a preferred embodiment of the present
invention, referred to herein as a "finger pointer" embodiment, a position sensing system includes a position sensing
transducer comprising a touch-sensitive surface disposed on
a substrate, such as a printed circuit board, including a
matrix of conductive lines. A first set of conductive lines
runs in a first direction and is insulated from a second set of
conductive lines running in a second direction generally
perpendicular to the first direction. An insulating layer is
disposed over the first and second sets of conductive lines.
The insulating layer is thin enough to promote significant
capacitive coupling between a finger placed on its surface
and the first and second sets of conductive lines.
Sensing electronics respond to the proximity of a finger to
translate the capacitance changes of the conductors caused
by finger proximity into position and touch pressure infor-

5,495,077
5

6

mation. Its output is a simple X, Y and pressure value of the
FIG. Ie is a composite view of the object position sensor
transducer of FIGS. la and lb showing both the top and
one object on its surface.
bottom conductive trace layers.
Different prior art pad scan techniques have different
FIG. ld is a cross-sectional view of the object position
advantages in different environments. Parallel drive/sense
techniques according to the present invention allow input 5 sensor transducer of FIGS. la-Ie.
samples to be taken simultaneously, thus all channels are
FIG. 2 is a block diagram of sensor decoding electronics
affected by the same phase of an interfering electrical signal,
which may be used with the sensor transducer in accordance
greatly simplifying the signal processing and noise filtering.
with a preferred embodiment of the present invention.
There are two drive/sense methods employed in the touch
FIG. 3a is a simplified schematic diagram of a charge
sensing technology of the present invention. According to a 10 integrator circuit which may be used in the present invenfirst and presently preferred embodiment of the invention,
tion.
the voltages on all of the X lines of the sensor matrix are
FIG. 3b is a schematic diagram of an illustrative schesimultaneously moved, while the voltages of the Y lines are
matic diagram of the charge integrator circuit of FIG. 3a.
held at a constant voltage, with the complete set of sampled
FIG. 4 is a timing of the operation of charge integrator
points simultaneously giving a profile of the finger in the X 15
circuit of FIGS. 3a and 3b.
dimension. Next, the voltages on all of the Y lines of the
FIG. 5 is a schematic diagram of an illustrative filter and
sensor matrix are simultaneously moved, while the voltages
sample/hold circuit for use in the present invention.
of the X lines are held at a constant voltage to obtain
FIG. 6a is a schematic diagram of an illustrative minimum
complete set of sampled points simultaneously giving a
20 selector and subtractor circuit including peak rejection
profile of the finger in the other dimension.
which may be employed in the present invention, showing
According to a second drive/sense method, the voltages
circuit details of four individual channels and their interon all of the X lines of the sensor matrix are simultaneously
connection.
moved in a positive direction, while the voltages of the Y
FIG. 6b is a representation of what the output of the
lines are moved in a negative direction. Next, the voltages on 25
minimum selector and subtractor circuit of FIG. 6a would be
all of the X lines of the sensor matrix are simultaneously
like without the background level removed.
moved in a negative direction, while the voltages of the Y
lines are moved in a positive direction. This technique
FIG. 6e is a representation of the output of the minimum
doubles the effect of any transcapacitance between the two
selector and subtractor circuit of FIG. 6a with the backdimensions, or conversely, halves the effect of any parasitic 30 ground level removed.
capacitance to ground. In both methods, the capacitive
FIG. 7 is a schematic diagram of an illustrative OTA
information from the sensing process provides a profile of
circuit used in the minimum selector and subtractor circuit,
the proximity of the finger to the sensor in each dimension.
showing how the outputs Pout and Zout are derived, and
Both embodiments then take these profiles and calculate
further showing a current sink and source options, Pouto and
the centroid for X and Y position and integrate under the 35 Poutp, respectively, for the Pout output.
curve for the Z pressure information. The position sensor of
FIG. 8 is a schematic diagram of an illustrative maximum
these embodiments can only report the position of one object
detector circuit which may be used in the present invention.
on its sensor surface. If more than one object is present, the
FIG. 9a is a schematic diagram of an illustrative position
position sensor of this embodiment computes the centroid
encoder circuit which may be used in the present invention.
position of the combined set of objects. However, unlike 40
FIG. 9b is a schematic diagram of an P-type OTA circuit
prior art, because the entire pad is being profiled, enough
which may be used in the position encoder circuit of the
information is available to discern simple multi-finger gespresent invention.
tures to allow for a more powerful user interface.
FIG. ge is a schematic diagram of an N-type OTA circuit
According to another aspect of the present invention,
several power reduction techniques which can shut down the 45 which may be used in the position encoder circuit of the
present invention.
circuit between measurements have been integrated into the
FIG. 10 is a schematic diagram of an illustrative ZSum
system. This is possible because the parallel measurement
circuit which may be used in the present invention.
technique according to the present invention is so much
faster than prior art techniques.
DETAILED DESCRIPTION OF A PREFERRED
According to a further aspect of the invention, noise 50
EMBODIMENT
reduction techniques that are focused on reducing noise
produced in typical computer environments are integrated
This application is a continuation-in-part of co-pending
into the system.
application Ser. No. 07/895,934, filed Jun. 8, 1992. The
According to yet another aspect of the present invention,
present invention continues the approach disclosed in the
a capacitance measurement technique which is easier to 55 parent application and provides more unique features not
calibrate and implement is employed.
previously available. These improvements provide increased
sensitivity, and greater noise rejection, increased data acquiBRIEF DESCRIPTION OF THE DRAWINGS
sition rate and decreased power consumption.
Those of ordinary skill in the art will realize that the
FIG. la is a top view of an object position sensor 60
following description of the present invention is illustrative
transducer according to a presently preferred embodiment of
only and not in any way limiting. Other embodiments of the
the invention showing the object position sensor surface
invention will readily suggest themselves to such skilled
layer including a top conductive trace layer and conductive
persons.
pads connected to a bottom trace layer.
The present invention brings together in combination a
FIG. lb is a bottom view of the object position sensor 65
number of unique features which allow for new applications
transducer of FIG. 1 a showing the bottom conductive trace
not before possible. Because the object position sensor of the
layer.

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present invention has very low power requirements, it is
beneficial for use in battery operated or low power applications such as lap top or portable computers. It is also a very
low cost solution, has no moving parts (and is therefore
virtually maintenance free), and uses the existing printed
circuit board traces for sensors. The sensing technology of
the present invention can be integrated into a computer
motherboard to even further lower its cost in computer
applications. Similarly, in other applications the sensor can
be part of an already existent circuit board.
Because of its small size and low profile, the sensor
technology of the present invention is useful in lap top or
portable applications where volume is important consideration. The sensor technology of the present invention
requires circuit board space for only a single sensorinterface
chip that can interface directly to a microprocessor, plus the
area needed on the printed circuit board for sensing.
The sensor material can be anything that allows creation
of a conductive XIY matrix of pads. This includes not only
standard PC board, but also flexible PC board, conductive
elastomer materials, silk-screened conductive lines, and
piez-oelectric Kynar plastic materials. This renders it useful
as well in any portable equipment application or in human
interface where the sensor needs to be molded to fit within
the hand.
The sensor can be conformed to any three dimensional
surface. Copper can be plated in two layers on most any
surface contour producing the sensor. This will allow the
sensor to be adapted to the best ergonomic form needed for
a application. This coupled with the "light-touch" feature
will make it effortless to use in many applications. The
sensor can also be used in an indirect manner, i.e it can have
a conductive foam over the surface and be used to detect any
object (not just conductive) that presses against it's surface.
Small sensor areas are practical, Le., a presently conceived embodiment takes about 1.5"x1.5" of area, however
those of ordinary skill in the art will recognize that the area
is scalable for different applications. The matrix area is
scaleable by either varying the matrix trace spacing or by
varying the number of traces. Large sensor areas are practical where more information is needed.
Besides simple X and Y position information, the sensor
technology of the present invention also provides finger
pressure information. This additional dimension of information may be used by programs to control special features
such as "brush-width" modes in Paint programs, special
menu accesses, etc., allowing provision of a more natural
sensory input to computers. It has also been found useful for
implementing "mouse click and drag" modes and for simple
input gestures.
The user will not even have to touch the surface to
generate the minimum reaction. This feature can greatly
minimize user strain and allow for more flexible use.
The sense system of the present invention depends on a
transducer device capable of providing position and pressure
information regarding the object contacting the transducer.
Referring first to FIGS. 1a-1d, top, bottom, composite, and
cross-sectional views, respectively, are shown of a presently-preferred touch sensor array for use in the present
invention. Since capacitance is exploited by this embodiment of the present invention, the sensor surface is designed
to maximize the capacitive coupling.
A presently preferred sensor array 10 according to the
present invention comprises a substrate 12 including a set of
first conductive traces 14 disposed on a top surface 16
thereof and run in a first direction to comprise row positions

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of the array. A set of second conductive traces 18 are
disposed on a bottom surface 20 thereof and run in a second
direction preferably orthogonal to the first direction to form
the column positions of the array. The sets of first and second
conductive traces 14 and 18 are alternately in contact with
periodic sense pads 22 comprising enlarged areas, shown as
diamonds in FIGS. la-Ie. While sense pads 22 are shown
as diamonds in FIGS. la-Ie, any shape, such as circles,
which allows close packing of the sense pads, is equivalent
for purposes of this invention. As an arbitrary convention
herein, the set of first conductive traces 14 will be referred
to as being oriented in the "X" or "row" direction and may
be referred to herein sometimes as "X lines" and the set of
second conductive traces 18 will be referred to as being
oriented in the "Y" or "column" direction and may be
referred to herein sometimes as "Y lines".
The number and spacing of these sense pads 22 depends
upon the resolution desired. For example, in an actual
embodiment constructed according to the principles of the
present invention, a 0.10 inch center-to-center diamondshaped pattern of sense pads disposed along a matrix of 15
rows and 15 columns of conductors is employed. Every
other sense pad 22 in each direction in the pad pattern is
connected sets of first and second conductive traces 14 and
18 on the top and bottom surfaces 16 and 20, respectively of
substrate 12.
Substrate 12 may be a printed circuit board, a flexible
circuit board or any of a number of available circuit interconnect technology structures. Its thickness is unimportant
as long as contact may be made therethrough from the
bottom conductive traces 18 to their sense pads 22 on the top
surface 16. The printed circuit board comprising substrate 12
can be constructed using standard industry techniques.
Board thickness is not important. Connections from the
sense pads 22 to the bottom traces 18 may be made employing standard plated-through hole techniques well known in
the printed circuit board art.
In an alternate embodiment of the present invention, the
substrate material 12 may have a thickness on the order of
0.005 to 0.010 inches. Then the diamonds on the top surface
16 and the plated thru holes that connect to the bottom
surface traces 18, can be omitted, further reducing the cost
of the system.
An insulating layer 24 is disposed over the sense pads 22
on top surface 16 to insulate a human finger or other object
therefrom. Insulating layer 24 is preferably a thin layer (Le.,
approximately 5 mils) to keep capacitive coupling large and
may comprise a material, such as mylar, chosen for its
protective and ergonomic characteristics. The term "significant capacitive coupling" as used herein shall mean capacitive coupling having a magnitude greater than about 0.5 pF.
There are two different capacitive effects taking place
when a finger approaches the sensor array 10. The first
capacitive effect is trans-capacitance, or coupling between
sense pads 22, and the second capacitive effect is selfcapacitance, or coupling to virtual ground. Sensing circuitry
is coupled to the sensor array 10 ofthe present invention and
responds to changes in either or both of these capacitances.
This is important because the relative sizes of the two
capacitances change greatly depending on the user environment. The ability of the present invention to detect changes
in both self capacitance and trans-capacitance results in a
very versatile system having a wide range of applications.
According to the preferred embodiment of the invention,
a position sensor system including sensor array 10 and
associated touch detector circuitry will detect a finger posi-

5,495,077
9
tion on a matrix of printed circuit board traces via the
capacitive effect of finger proximity to the sensor array 10.
The position sensor system will report the X, Y position of
a finger placed near the sensor array 10 to much finer
resolution than the spacing between the row and column 5
traces 14 and 18. The position sensor according to this
embodiment of the invention will also report a Z value
proportional to the outline of that finger and hence indicative
of the pressure with which the finger contacts the surface of
insulating layer 24 over the sensor array
10
According to the presently preferred embodiment of the
invention, a very sensitive, light-touch detector circuit may
be provided using adaptive analog VLSI techniques. The
circuit of the present invention is very robust and calibrates
out process and systematic errors. The detector circuit of the 15
present invention will process the capacitive input information and provide digital information to a microprocessor.
According to this embodiment of the invention, sensing
circuitry is contained on a single sensor processor integrated
circuit chip. The sensor processor chip can have any number 20
of X and Y "matrix" inputs. The number of X and Y inputs
does not have to be equal. The Integrated circuit has a digital
bus as output. In the illustrative example disclosed in FIGS.
1a-1d herein, the sensor array 10 has 15 traces in both the
Y and Y directions. The sensor processor chip thus has 15 X 25
inputs and 15 Y inputs.
The X and Y matrix nodes are driven and sensed in
parallel, with the capacitive information from each line
indicating how close a finger is to that node. The scanned
information provides a profile of the finger proximity in each 30
dimension. According to this aspect of the present invention,
the profile centroid is derived in both the X and Y directions
and is the position in that dimension. The profile curve of
proximity is also integrated to provide the Z information.
35
There are two drive and sense methods employed in the
touch sensing technology of the present invention. According to a first and presently preferred embodiment of the
invention, the voltages on all of the X lines of the sensor
array 10 are simultaneously moved, while the voltages of the
40
Y lines are held at a constant voltage. Next, the voltages on
all of the Y lines of the sensor array 10 are simultaneously
moved, while the voltages of the X lines are held at a
constant voltage. This scanning method accentuates the
measurement of capacitance to virtual ground provided by 45
the finger. Those of ordinary skill in the art will recognize
that order of these two steps is somewhat arbitrary and may
be reversed.
According to a second drive/sense method, the voltages
on all of the X lines of the sensor array 10 are simultaneously 50
moved in a positive direction, while the voltages of the Y
lines are moved in a negative direction. Next, the voltages on
all of the X lines of the sensor array 10 are simultaneously
moved in a negative direction, while the voltages of the Y
lines are moved in a positive direction. This second drive/55
sense method accentuates transcapacitance and de-emphasizes virtual ground capacitance. As with the first drivel
sense method, those of ordinary skill ill the art will recognize
that order of these two steps is somewhat arbitrary and may
60
be reversed.

10

sions do not need to be orthogonal. For example, they can be
radial or of any other nature to match the contour of the
sensing pad and the needs of the system.
The capacitance at each sensor matrix node is measured
simultaneously using charge integrator circuits 32-1 through
32-n. The function of each charge integrator is to develop an
output voltage proportional to the capacitance sensed on the
corresponding X matrix line.
According to the presently preferred drive/sense method,
the capacitance measurements are performed simultaneously across all inputs in one dimension to overcome a
problem which is inherent in all prior art approaches that
scan individual inputs. The problem with the prior-art
approach is that it is sensitive to high frequency and large
amplitude noise (large dv/dt noise) that is coupled to the
circuit via the touching object. Such noise may distort the
finger profile because of noise appearing in a later scan cycle
but not an earlier one, due to a change in the noise level. The
present invention overcomes this problem by taking a snapshot of all inputs simultaneously. The injected noise is
proportional to the finger signal strength across all inputs
and therefore symmetric around the finger centroid. Because
it is symmetric around the finger centroid it does not affect
the finger position.
Because of the nature of the charge integrator circuits
32-1 through 32-n, their outputs will be changing over time
and will have the desired voltage output for only a short
time. This desired voltage is captured by the filter and
sample/hold circuits 34-1 through 34-n. As controlled by the
control circuitry, 36, the filter and sample/hold circuits 34-1
through 34-n will capture the desired voltage and store it.
Additionally, the result may also be filtered depending on the
size of the sample and hold capacitor in the cell.
The filter and sample/hold circuits 34-1 through 34-n then
provides an input for the Minimum Selector and Subtractor
circuit 38, which computes an average of its n smallest input
values (n=3 is presently preferred) and subtracts that value
from each input. Minimum Selector and Subtractor circuit
38 then generates a current output for every input which is
proportional to the difference at that input between the actual
value at the input and the computed average minimum value.
This circuit performs the task of subtracting out the background capacitance seen by the sensing circuitry and then
providing a current proportional to the additional capacitance seen above the background level. If n=l, then the
minimum value is selected. Any n> 1 will select an average
of the n values.
This current is then replicated and sent to two destinations. One copy is sent to the position encoder circuit 40, and
the second is sent to the Zsum circuit 42.
The position encoder circuit 40 uses the current inputs as
weights, and provides a scaled weighted mean (centroid) of
the set of input currents and their relation to their position in
the sensor. Position encoder circuit 40 is a linear position
encoder having a voltage output which varies between the
power supply rails. Because the circuit produces a continuous weighted mean over all input values, it is capable of
interpolation to a much finer resolution than the spacing of
the matrix grid spacing.
The output of the position encoder circuit 40 is then
Referring now to FIG. 2, a block diagram of the presently
presented to sample/hold circuit 44-1 and analog to digital
preferred sensing circuitry 30 for use according to the
(ND) converter 46-1. The operation of this portion of the
present invention is presented. This block diagram shows the
circuit uses devices that are well known to those knowlsensing circuitry 30 in one dimension (X) only. Those of
ordinary skill in the art will appreciate that an identical 65 edgeable in the art.
The minimum selector and subtractor circuit 38 also
circuit would be used for sensing the opposite (Y) dimension. Such skilled persons will further note that these dimengenerates a second set of outputs which are all tied together

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and sent to the Zsum circuit 42. Since these lines are shorted
together, the individual output currents are all summed
together. The total effect of the finger on the sensor in one
dimension is thus integrated, producing a current sum result
proportional to the pressure or proximity of the input object.
The Zsum circuit 42 takes this current sum and then converts
it back to a voltage which is proportional to the current sum.
Those of ordinary skill in the art will appreciate that there
are many conversion choices depending on the particular use
to which the invention is put, it can be a linear conversion,
square root (compressive), or squared (expansive). In presently preferred embodiment a compressive conversion is
chosen to compress large changes and emphasize small
changes, since detecting very light touches is of particular
interest.
The output of the ZSum circuit 42 is presented to a
samplelhold circuit 44-2 which stores the results. The output
of sample hold circuit 44-2 drives AID converter circuit 46-2
which converts the analog information to a digital form
useable by microcomputers.
Control circuitry 36 of FIG. 2 orchestrates the operation
of the remainder of the circuitry. Because the system is
discretely sampled and pipelined in its operation, control
circuitry 36 is present to manage the signal flow. The
functions performed by control circuitry 36 may be conventionally developed via what is commonly known in the art
as a state machine or by a microcontroller.
The output of the filter and sample/hold circuits, 34-1 thru
34-n, is also monitored by a maximum detector circuit 47.
The purpose of this section of circuitry is to generate an
interrupt signal to a microprocessor if there is a finger signal
greater than a preset threshold. The maximum detector
circuit 47 outputs a signal which is related to the largest
input voltage. This signal is then compared against a predetermined threshold, noted as VTHMAX with the comparator 48. If the signal is greater than the preset threshold
the comparator will output a logic I level which, after
conditioned with proper timing thru AND gate 49, provides
an interrupt signal to a microprocessor. Those skilled in the
art will recognize that this signal is not limited to being an
interrupt and could be used for polling, for example, or in
other ways that better fit the needs of the entire system.
The structure and operation of the individual blocks of
FIG. 2 will now be disclosed. Referring now to FIGS. 3a, 3b,
and 4, a typical charge integrator circuit will be described.
Charge integrator circuit 32 is shown as a simplified schematic diagram in FIG. 3a and as an illustrative schematic
diagram in FIG. 3b. The timing of the operation of charge
integrator circuit 32 is shown in FIG. 4.
Charge integrator circuit 32 is based on the fundamental
physical phenomena of using a current to charge a capacitor.
If the capacitor is charged for a constant time by a constant
current then a voltage will be produced on the capacitor
which is inversely proportional to the capacitance. The
capacitance to be charged is the sensor array line capacitance
in parallel with an internal capacitor. This internal capacitor
will contain the voltage of interest.
Referring now to FIG. 3a, a simplified schematic diagram
of an illustrative charge integrator circuit 32 is shown. A
charge integrator circuit input node 50 is connected to one
of the X (or Y) lines of the sensor array. A first shorting
switch 52 is connected between the charge integrator circuit
input node 50 and VDD' the positive supply rail. A second
shorting switch 54 is connected between the charge integrator circuit input node 50 and ground, the negative supply
rail. A positive constant current source 56 is connected to

VDD' the positive supply rail and to the charge integrator
circuit input node 50 and through a first current source
switch 58. A negative constant current source 60 is connected to ground and to the charge integrator circuit input
node 50 and through a second current source switch 62.
A first internal capacitor 64 is connected between VDD
and output node 66 of charge integrator circuit 32. A positive
voltage storage switch 68 is connected between output node
66 and input node 50. A second internal capacitor 70 has one
of its plates connected to ground through a switch 72 and to
output node 66 of charge integrator circuit 32 through a
switch 74, and the other one of its plates connected to input
node 50 through a negative voltage storage switch 76 and to
VDD through a switch 78. The capacitance of first and
second internal capacitors 64 and 70 should be a small
fraction (i.e., about 10%) of the capacitance of the individual
sensor array lines. In a typical embodiment, the sensor array
line capacitance will be about 10 pF and the capacitance of
capacitors 64 and 70 should be about I pF.
According to the presently preferred embodiment of the
invention, the approach used is a differential measurement
for added noise immunity, the benefit of which is that any
low frequency common mode noise gets subtracted out. For
the following discussion, it is to be assumed that all switches
are open unless they are noted as closed. First, the sensor
array line is momentarily shoaled to VDD through switch 52,
switch 68 is closed connecting capacitor 64 in parallel with
the capacitance of the sensor line. Then the parallel capacitor
combination is discharged with a constant current from
current source 60 through switch 62 for a fixed time period.
At the end of the fixed time period, switch 68 is opened, thus
storing the voltage on the sensor array line on capacitor 64.
The sensor line is then momentarily shorted to ground
through switch 54, and switches 72 and 76 are closed to
place capacitor 70 in parallel with the capacitance of the
sensor line. Switch 58 is closed and the parallel capacitor
combination is charged with a constant current from current
source 56 for a fixed time period equal to the fixed time
period of the first cycle. At the end of the fixed time period,
switch 76 is opened, thus storing the voltage on the sensor
array line on capacitor 70.
The first and second measured voltages are then averaged.
This is accomplished by opening switch 72 and closing
switches 78 and 74, which places capacitor 70 in parallel
with capacitor 64. Because capacitors 64 and 70 have the
same capacitance, the resulting voltage across them is equal
to the average of the voltages across each individually. This
final result is the value that is then passed onto the appropriate one of filter and samplelhold circuits 34-1 through
34-n.
The low frequency noise, notably 50/60 Hz and their
harmonics, behaves as a DC current component that adds in
one measurement and subtracts in the other. When the two
results are added together that noise component is canceled
out. The amount of noise rejection is a function of how
quickly in succession the two charge/discharge cycles are
performed. One of the reasons for the choice of this charge
integrator circuit is that it allows measurements to be taken
quickly.
Referring now to FIG. 3b, a more complete schematic
diagram of an illustrative embodiment of charge integrator
circuit 32 of the simplified diagram of FIG. 3a is shown.
Input node 50 is shown connected to VDD and ground
through pass gates 80 and 82, which replace switches 52 and
54 of FIG. 3a. Pass gate 80 is controlled by a signal ResetUp
presented to its control input and pass gate 82 is controlled

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by a signal ResetDn presented to its control input. Those of
ordinary skill in the art will recognize that pass gates 80 and
82, as well as all of the other pass gates which are represented by the same symbol in FIG. 3b may be conventional
CMOS pass gates as are known in the art. The convention
used herein is that the pass gate will be off when its control
input is held low and will be on and present a low impedance
connection when its control input is held high.
P-Channel MOS transistors 84 and 86 are configured as a
current mirror. P-Channel MOS transistor 84 serves as the
current source 56 and pass gate 88 serves as switch 58 of
FIG. 3a. The control input of pass gate 88 is controlled by
a signal StepUp,
N-Channel MOS transistors 92 and 94 are also configured
as a current mirror. N-Channel MOS transistor 92 serves as
the current source 60 and pass gate 96 serves as switch 62
of FIG. 3a. The control input of pass gate 96 is controlled by
a signal StepDn. P-Channel MOS transistor 90 and N-Channel MOS transistor 98 are placed in series with P-Channel
MOS current mirror transistor 86 and N-Channel MOS
current mirror transistor 94. The control gate of P-Channel
MOS transistor 90 is driven by an enable signal EN, which
turns on P-Channel MOS transistor 90 to energize the
current mirrors. This device is used as a power conservation
device so that the charge integrator circuit 32 may be turned
off to conserve power when it is not in use.
N-Channel MOS transistor 98 has its gate driven by a
reference voltage Vref, which sets the current through current mirror transistors 86 and 94. The voltage Vref may be
individually adjusted for each charge integrator circuit 32 to
compensate for manufacturing variations. Each Vref may be
developed from a analog programmable voltage source,
such as described in van Steenwijk, Hoen, and Wallinga "A
Nonvolatile Analog Voltage Programmable Voltage Source
Using VIP MOS EEPROM Structure," IEEE Journal of
Solid State Circuits, Jul. 1993. Alternatively, a writable
analog reference voltage storage device, such as disclosed in
U.S. Pat. No. 5,166,562 may be employed. This allows the
circuit to be calibrated in the factory to zero out process
variations as well as capacitance variations in the sensor.
The calibration focus is to generate a constant and equal
output from all the charge integrator circuits 32-1 through
32-n of FIG. 2 if no finger is present. Although the present
approach is very robust, those of ordinary skill in the art will
also appreciate an embodiment in which calibration will be
allowed to occur in real time (via long time constant
feedback) thereby zeroing out any long term effects due to
sensor environmental changes.
Note that proper sizing ofMOS transistors 94 and 98 may
provide temperature compensation. This is accomplished by
taking advantage of the fact that the threshold ofN-Channel
MOS transistor 98 reduces with temperature while the
mobility of both N-Channel MOS transistors 94 and 98
reduce with temperature. The threshold reduction has the
effect of increasing the current while the mobility reduction
has the effect of decreasing the current. By proper device
sizing these effects can cancel each other out over a significant part of the operating range.
Capacitor 64 has one plate connected to VDD and the other
plate connected to the output node 66 and to the input node
50 through pass gate 100, shown as switch 68 in FIG. 3a.
The control input of pass gate 100 is driven by the control
signal SUp. One plate of capacitor 70 is connected to input
node 50 through pass gate 102 (switch 76 in FIG. 3a) and to
VDD through pass gate 104 (switch 72 in FIG. 3a). The
control input of pass gate 102 is driven by the control signal

SDn and the control input of pass gate 104 is driven by the
control signal ChUp. The other plate of capacitor 70 is
connected to ground through N-Channel MOS transistor 106
(switch 72 in FIG. 3a) and to output node 66 through pass
gate 108. The control input of pass gate 108 is driven by
control signal Share.
Referring now to FIGS. 3a, 3b and the timing'diagram of
FIG. 4; the operation of charge integrator circuit 32 during
one scan cycle may be observed. First the EN (enable)
control signal goes active by going to 0 v. This turns on the
current mirrors and energizes the charge and discharge
current sources, MOS transistors 84 and 92. The ResetUp
control signal is active high at this time, which shorts the
input node 50 (and the sensor line to which it is connected)
to VDD' The SUp control signal is also active high at this
time which connects capacitor 64 and the output node 66 to
input node 50. This arrangement guarantees that the following discharge portion of the operating cycle always starts
from a known equilibrium state.
The discharge process starts after ResetUp control signal
goes inactive. The StepDn control signal goes active, connecting MOS transistor 92, the discharge current source, to
the input node 50 and its associated sensor line. StepDn is
active for a set amount of time, allowing the combined
capacitance of the sensor line and capacitor 64 to charge
down during that time. StepDn is then turned off. A short
time later the SUp control signal goes inactive, storing the
measured voltage on capacitor 64 to end the discharge cycle.
Next, ResetDn control signal becomes active and shorts
the sensor line to ground. Simultaneously the SDn and ChDn
control signals become active and connect capacitor 70
between ground and the sensor line. Capacitor 70 is discharged to ground, guaranteeing that the following charge
up cycle always starts from a known state.
The chargeup cycle starts after ResetDn control signal
becomes inactive and the StepUp control signal becomes
active. At this point the current charging source, MOS
transistor 84, is connected to the sensor line and the sensor
line charges up. The StepUp control signal is active for a set
amount of time (preferably equal to the time for the previously mentioned cycle) allowing the capacitance to charge,
and then it is turned off. The SDn control signal then goes
inactive, leaving the measured voltage across capacitor 70.
The averaging cycle now starts. First the voltage on
capacitor 70 is level shifted. This is done by the ChDn
control signal going inactive, letting one plate of the capacitor 70 float. Then the ChUp control signal goes active,
connecting the second plate of the capacitor ter V DD' Then
the Share control signal becomes active which connects the
first plate of capacitor 70 to output node 66, thus placing
capacitors 64 and 70 in parallel. This has the effect of
averaging the voltages across the two capacitors, thus subtracting out common mode noise as previously described.
This average voltage is also then available on output node
66.
According to the present invention, two different drive!
sense methods have been disclosed. Those of ordinary skill
in the art will readily observe that the charge integrator
circuit 32 disclosed with reference to FIGS. 3a, 3b, and 4 is
adaptable to operate according to either scanning method
disclosed herein.
As is clear from an understanding of the operation of
charge integrator circuit 32, its output voltage is orily
available for a short period of time. In order to capture this
voltage a sample/hold circuit is used. Referring now to FIG.
5, a schematic diagram of an illustrative filter and sample!

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hold circuit is presented. Those of ordinary skill in the art
will recognize this circuit, which comprises an input node
112, a pass gate 114 having a control input driven by a
Sample control signal, a capacitor 116 connected between
the output of the pass gate 114 and a fixed voltage such as
ground, and an output node comprising the common connection between the capacitor 116 and the output of the pass
gate 114. In a typical embodiment, capacitor 116 will have
a capacitance of about 10 pF.
The sample/hold circuit of FIG. 5 is well known in the art,
but is applied in a way so that it acts as a filter as well. The
filter time constant is K times the sample signal period,
where K is the ratio of capacitor 116 to the sum of capacitors
64 and 70 of the charge integrator circuit 32 of FIGS. 3a and
3b. This filter further reduces noise injection. In the preferred embodiment, K=1O/2=5.
As shown in FIG. 2, the output of all of the filter and
sample/hold circuits 34-1 thru 34-n drive minimum selector
and subtractor circuit 38. Referring now to FIG. 6a, a
schematic diagram of an illustrative minimum selector and
subtractor circuit 38 useful for employment in the present
invention is shown. The illustrative circuit of FIG. 6a is
shown having four channels, although those of ordinary skill
in the art will readily recognize that the circuit could be
arbitrarily extended to a larger number of channels.
The minimum selector and subtractor circuit 38 is
designed to take a set of inputs, detect the average of the
three smallest input values and subtract that average value
from each individual value in the entire input set. The circuit
then produces a current which is proportional to this subtracted value, which for most background inputs will be
zero. This sequence of steps is illustrated in FIGS. 6b and 6c
for an example where there are 15 inputs (Xl to XIS) in the
set. FIG. 6b shows the input to the minimum selector/
subtractor circuit as generated by the Filter circuits, 34-1
thru 34-n. The drawing shows a typical finger profile with
the background or minimum level noted. After the minimum
selector and subtractor circuit 38 processes the input, it
produces an output like that shown in FIG. 6c, which is the
input set with the background value subtracted out.
Each individual channel, even though constituting a
single functional unit, can be thought of as consisting of a
minimum selector circuit 120 and a subtractor circuit 122. In
the minimum selector circuit 120, the active elements are
P-channel MOS transistors 124a-124d, each having its
source electrode connected to an intermediate node
126a-126d, its gate electrode connected to the channel input
nodes 128a-128d, its drain electrode connected to the drain
electrodes of N-channel MOS current-limiting transistors
130a-130d, respectively. N-channel MOS transistors
130a-130d have their source electrodes connected to a fixed
voltage, such as ground, and their gate electrodes held at a
potential V sink above (more positive than) their source
electrodes such that they function as current sinks with
limited voltage compliance.
Each intermediate node 126a-126d is also connected to a
current source 132a-132d that supplies the operating current
of the P-channel MOS transistors 124a-124d from a fixed
voltage source V DD' Intermediate nodes 126a-126d are also
connected to the non-inverting input of an operational
transconductance amplifier 134a-134d (OTA), which comprises the heart of the subtractor circuit 122 of each channel.
A pass gate 136a-136d allows connecting the intermediate
node 126a-126d to a minimum rail 138 which is common to
all signal channels in the system. The inverting input of each
OTA 134a-134d is connected to a storage capacitor

140a-140d as well as to one end of a pass gate 142a-142d

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which permits selectively connecting the inverting input of
the OTA 134a-134d to its Pout output 144a-144d. Each
OTA also has a Zout output 146a-146d.
The N-Channel MOS current sink transistors 130a-130d
have the effect of limiting the current that anyone of the
P-channel MOS transistors 124a-124d can draw. As is well
known to those of ordinary skill in the art, this is because the
common drain connections of the N-channel and P-channel
MOS transistors will assume such a potential as to reduce
the drain-to-source voltage difference of one or the other of
the two transistors far enough to prevent it from drawing
more current than the other transistor allows to flow.
If the sink current of N-Channel MOS current sink
transistors 130a-130d is chosen to be larger than the current
of sources 132a-132d but smaller than the sum of all source
currents, then no single P-Channel MOS transistor
124a-124d can conduct all the current in the minimum
selection phase of operation. Instead, several transistors
have to share it. Thus it may be seen that the minimum select
and subtractor circuit of FIG. 6a selects not the absolute
minimum of the input voltages of all channels, but rather an
average of the several lowest input voltages from among all
channels. The gate voltage of the N-Channel MOS currentlimiting transistors l30a-130d is selected to set their saturation current to be WIn times the value of the current source
132, where W is the number of channels present in the
system and n is the number of channels to be averaged over
to obtain the minimum.
For example, assuming an embodiment including fifteen
identical signal channels, and the currents of the N-channel
MOS transistors 130 in each channel are five times larger
than the currents of current sources 132, then the three
P-channel transistors 124 with the lowest gate potentials
must share the total system current because the total is equal
to fifteen times the source current. The effect is to reject or
at least attenuate negative peaks on the input voltages. This
property is highly desirable in applications where it is
expected that several or most channel input voltages will
always be equal to a common minimum or baseline potential, but that considerable noise existing on the inputs may
create false negative peaks.
As may be seen from FIG. 6a, each OTA 134a-134d has
two output types, designated as Pout 144a-144d and Zout
146a-146d. Referring now to FIG. 7, the circuitry for
generating these outputs may be seen in detail. As shown in
FIG. 7, each OTA comprises N-Channel MOS input transistors 148 and 150, P-Channel MOS current mirror pairs
configured from transistors 152 and 154, 156 and 158, and
N-Channel MOS bias transistor 160. An N-Channel MOS
current mirror comprising transistors 162 and 164 is connected to P-Channel current mirror transistors 152 and 158
as shown.
As thus far described, the circuit is conventional and the
common drain node of transistors 158 and 164 would form
an output node for the circuit. An extra P-channel MOS
transistor 166 and N-Channel MOS transistor 168 are added
to the circuit to form a second output node by replicating the
output buffer of the typical wide range output transamplifier.
The common drain node of transistors 166 and 168, through
pass gate 170, forms the Pout output section of the circuit.
The diode-connected N-Channel MOS transistor 172
between the common drain node of transistors 158 and 164
and the Zout node assures that the Zout lines of minimum
selector and subtractor circuit 38 will only source current,
thus guaranteeing that only object-created signals above the
baseline contribute to the Zout signal.

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In the presently preferred embodiment of the invention,
the Pout outputs of half of the minimum selector and
subtractor circuit 38 are configured as current source outputs
and may be designated Poutp outputs. The other half of the
outputs are current sink outputs and may be designated
Poutn outputs. This feature is also shown in FIG. 7. Pass gate
170, controlled by control signal PosEn is present to disconnect the position encode load during the sample phase of
the minimum select and subtract circuit. The current source
node for the Poutp outputs is at the output of pass gate 170.
In this configuration, transistors 174 and 176 are not present
and Poutp goes to output Pout. This output would be used to
drive the BiasIn line of the position encoder OTA as shown
in FIG. 9b. The current sink path is developed from the
transamplifier output current at the output of pass gate 170.
That current is fed into an NMOS current mirror comprising
N-Channel MOS transistors 174 and 176. The drain of
N-Channel transistor 176 is the current sink Poutn output
node and is connected to output Pout. This output would be
used to drive the BiasIn line of the position encoder OTA as
shown in FIG. 9c.
Referring now to FIG. 8, an illustrative maximum detector circuit 47 for use in the present invention is shown. As
previously mentioned, the function of maximum detector
circuit 47 is to monitor the outputs of filter and samplelhold
circuits, 34-1 thru 34-n and to generate an interrupt signal to
a microprocessor if there a finger signal greater than a preset
threshold VTHMAX is present. Those skilled in the art will
recognize that the signal is not limited to being an interrupt
and could be used for other purposes such as polling etc.
Maximum detector circuit 47 includes an N-channel MOS
bias transistor 182 having its source connected to ground
and its gate connected to a bias voltage VBIAS at node 184.
The inputs of the maximum detector circuit 47 are connected
to the outputs of filter and samplelhold circuits 34-1 to 34-n
as shown in FIG. 2. In the maximum detector circuit 47
illustrated in FIG. 8, there are (n) inputs. Each input section
comprises a series pair of MOS transistors connected
between the drain ofN-channel MOS bias transistor 182 and
a voltage source VDD'
Thus, the input section for In l comprises P-channel MOS
current-limiting transistor 186 having its source connected
to VDD and its drain connected to the drain of N-channel
MOS input transistor 188. The gate ofN-channel MOS input
transistor 188 is connected to In l input node 190 and the gate
of P-channel MOS current-limiting transistor 186 is connected to a source of bias voltage VLBIAS at node 192.
Similarly, the input section for In2 comprises P-channel
MOS current-limiting transistor 194 having its source connected to VDD and its drain connected to the drain of
N-channel MOS input transistor 196. The gate ofN-channel
MOS input transistor 196 is connected to In2 input node 198
and the gate of P-channel MOS current-limiting transistor
194 is connected to node 192.
The input section for In3 comprises P-channel MOS
current-limiting transistor 200 having its source connected
to VDD and its drain connected to the drain of N-channel
MOS input transistor 202. The gate ofN-channel MOS input
transistor 202 is connected to In3 input node 204 and the gate
of P-channel MOS current-limiting transistor 200 is connected to node 192.
The input section for In(n) comprises P-channel MOS
current-limiting transistor 206 having its source connected
to VDD and its drain connected to the drain of N-channel
MOS input transistor 208. The gate ofN-channel MOS input
transistor 208 is connected to In(n) input node 210 and the

18
gate of N-channel MOS current-limiting transistor 206 is
connected to node 192. The sources of N-channel MOS
input transistors 188, 196, 202, and 208 are connected
together to the drain of N-channel MOS bias transistor 182.
5 The output of maximum detector circuit 47 is node 212 at
the common connection of the drain ofN-channel MOS bias
transistor 182 and the sources of the N-channel MOS input
transistors 188, 196, 202 and 208.
The maximum detector circuit 47 acts analogously to the
10 minimum detector circuit 46 of parent application Ser. No.
07/895,934, filed Jun. 8, 1992. The difference is that an
N-channel bias transistor is used instead of a P-channel bias
transistor and an N-channel transconductance amplifier is
used in place of a P-channel transconductance amplifier. The
15 result is the output will now track approximately an N-channel bias drop below the largest input (in non-averaging
mode), since that much difference is needed to guarantee at
least one input pair is on (l861l88, 1941196 , ... 2061208).
However for this circuit the output is not used for feed20 back, but is instead used to drive a comparator 48 (FIG. 2)
which is set to trip if the input is greater than the voltage
VThmax' If tripped, a MAX INTERRUPT signal is generated.
The MAX INTERRUPT is used to "wake-up" a microprocessor and tell it that there is an object detected at the sensor.
25 The signal is prevented from appearing on the MAX
INTERRUPT line by AND gate 49 and a control signal from
control circuitry 36. The control signal only allows the
interrupt signal to pass after the circuit has settled completely. The control signal presented to AND gate 49 may be
30 a SAMPLE signal which may be generated, for example, by
the trailing edge of the SHARE signal shown in FIG. 4.
As may be seen from FIG. 2, the outputs of minimum
selector and subtractor circuit 38 are presented to position
encoder circuit 40. There are two identical position encoder
35 circuits, one each for the X and Y directions. The function
of position encode circuit 40 is to convert the input information into a signal representing object proximity in the X
(or Y) dimension of the sensor array matrix. According to a
presently preferred embodiment of the invention, this circuit
40 will provide a scaled weighted mean (centroid) of the set of
input currents. The result is a circuit which is a linear
position encoder, having an output voltage which varies
between the power supply rails. Because it is a weighted
mean, it averages all current inputs and can in tum generate
45 an output voltage which represents an X (or Y) position with
a finer resolution than the spacing of the matrix grid spacing.
Referring now to FIG. 9a, a presently preferred embodiment of a position encoder circuit 40 of FIG. 2 is shown in
50 schematic diagram form. Because the position encoder circuits in the X and Y dimensions are identical, only one will
be shown. The position encoder circuit 40 of FIG. 9a is
shown having six inputs, but those of ordinary skill in the art
will recognize that, due to its symmetry, it may be arbitrarily
expanded for other numbers of inputs.
55
As presently preferred, position encoder circuit 38
includes a plurality of transconductance amplifiers 220-1
through 220-6 connected as followers. The outputs of all
amplifiers 220-1 through 220-6 are connected together to a
60 common node 222, which comprises the position encoder
output node of the circuit 38.
The non-inverting inputs of amplifiers 220-1 through
220-6 are connected to a resistive voltage divider network
comprising resistors 224, 226, 228, 230, 232, 234, and 236,
65 shown connected between VDD and ground.
Amplifiers 220-1 through 220-3 have P-channel MOS
bias transistors and differential pair inputs due to the input

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operating range between zero volts and V DIl2, and are
shown in schematic diagram form in FIG. 9b. The P-Channel MOS devices 250 and 252 form the differential input pair
while 254 and 256 form a current mirror load. This is the
standard configuration for a typical transconductance amplifier. Normally the bias current is provided by a P-Channel
MOS current source device at node 258. However in this
application the bias current is provided externally by the
Poutp output (output of pass gate 170 in FIG. 7.) of the
minimum selectorlsubtractor circuit through nodes 238, 240
and 242 (lIN! thru I lN3 , respectively, of FIG. 9a).
Amplifiers 220-4 through 220-6 have N-channel MOS
bias transistors and differential pair inputs due to the input
operating range between V Dd2 and V DD' and are shown in
schematic diagram form in FIG. 9c. The N-Channel MOS
transistors 260 and 262 form the differential input pair while
P-Channel MOS transistors 264 and 266 form a current
mirror load. This is the standard configuration for a typical
transconductance amplifier. Normally the bias current is
provided by aN-Channel MOS current source at node 268.
However in this application the bias current is provided
externally by the Poutn output (drain of transistor 174 in
FIG. 7) of the minimum selector and subtractor circuit
through nodes 244, 246 and 248 (11N4 thru IIN6' respectively,
of FIG. 9a). Those of ordinary skill in the art will readily
recognize that amplifiers 220-4 through 220-6 will be configured exactly like amplifiers 220-1 through 220-3, except
that all transistor and supply voltage polarities are reversed.
The position encoder circuit of FIG. 9a will provide a
weighted mean (centroid) of the input currents weighted by
the voltages on the resistor divider circuit to which the inputs
of the amplifiers 220-1 through 200-6 are connected. If the
resistors 224, 226, 228, 230, 232, 234, and 236 are all equal
then the result is a circuit which is a linear position encoder,
with its output voltage varying between the power supply
rails. Because it is a weighted mean, it averages all current
inputs which in tum generates an interpolated output. This
arrangement affords finer resolution than the voltage spacing
of voltage nodes "n" at the input. This is key to making a
dense circuit function. This circuit is an improvement of a
circuit described in DeWeerth, Stephen P., Analog VLSI
Circuits For Sensorimotor Feedback, Ph.D Thesis, California Institute of Technology, 1991.
The output voltage of X position encoder circuit 40 is
presented to samplelhold circuit 44-1, the output of which,
as is well known in the art, either follows the input or holds
a value present at the input depending on the state of its
control input. The structure and operation of samplelhold
circuits are well known in the art.
The output of samplelhold circuit 44-1 drives the input of
analog-to-digital (ND) converter 46-1. The output of AID
converter 46-1 is a digital value proportional to the position
of the object in the X dimension of the sensor array matrix
10.
Referring now to FIG. 10, a schematic diagram of a
presently preferred embodiment of a ZSum circuit 42 is
shown. ZSum circuit 42 takes a current as an input on node
270. If N-Channel MOS transistor 272 was not present then
the combination ofN-Channel MOS transistors 274 and 276
would be a current mirror and the current on input node 270
would appear on node 278. Transistor 272 is a source
degradation resistor that, depending on the gain setting on
node 280, reduces the current mirror transfer factor from an
ideal factor of 1 to something less than 1. The smaller the
voltage present on node 280 the smaller the transfer factor.
P-Channel MOS transistors 282 and 284 create another
current mirror that copies the current in node 278 into node

286, the output node. Diode connected N-Channel MOS
transistor 288 converts the current back into a voltage with
a square root transfer function or a compressive non-linearity. This is chosen to accentuate the low level currents and
hence is suited to process a light touch at the sensor.
The increased sensitivity of the touch sensor system of the
present invention allows for a lighter input finger touch
which makes it easy for human use. Increased sensitivity
also makes it easier to use other input objects, like pen styli,
etc. Additionally this sensitivity allows for a trade-off
against a thicker protective layer, or different materials,
which both allow for lower manufacturing costs.
Greater noise rejection allows for greater flexibility in use
and reduced sensitivity to spurious noise problems. Two
techniques are employed which allow derivation of the most
noise-rejection benefit.
Due to the drive and sense techniques employed in the
present invention, the data acquisition rate has been
increased by about a factor of 30 over the prior art. This
offers several obvious side effects. First, for the same level
of signal processing, the circuitry can be turned off most of
the time and reduce power consumption by roughly a factor
of 30 in the analog section of the design. Second, since more
data is available, more signal processing, such as filtering,
and gesture recognition, can be performed.
The sensor electronic circuit employed in the present
invention is very robust and calibrates out process and
systematic errors. It will process the capacitive information
from the sensor and provide digital information to an
external device, for example, a microprocessor.
Because of the unique physical features of the present
invention, there are several ergonomically interesting applications that were not previously possible. Presently a Mouse
or Trackball is not physically convenient to use on portable
computers. The present invention provides a very convenient and easy-to-use cursor position solution that replaces
those devices.
In mouse-type applications, the sensor of the present
invention may be placed in a convenient location, e.g.,
below the "space bar" key in a portable computer. When
placed in this location, the thumb of the user may be used as
the position pointer on the sensor to control the cursor
position on the computer screen. The cursor may then be
moved without the need for the user's fingers to leave the
keyboard. Ergonomically, this is similar to the concept of the
Macintosh Power Book with it's trackball, however the
present invention provides a significant advantage in size
over the track ball. Extensions of this basic idea are possible
in that two sensors could be placed below the "space bar"
key for even more feature control.
The computer display with it's cursor feedback is one
small example of a very general area of application where a
display could be a field of lights or LED's, a LCD display,
or a CRT. Examples include touch controls on laboratory
equipment where present equipment uses a knoblbuttonl
touch screen combination. Because of the articulating ability
of this interface, one or more of those inputs could be
combined into one of our inputs.
Consumer Electronic Equipment (stereos, graphic equalizers, mixers) applications often utilize significant front
panel surface area for slide potentiometers because variable
control is needed. The present invention can provide such
control in one small touch pad location. As Electronic Home
Systems become more common, denser and more powerful
human interface is needed. The sensor technology of the
present invention permits a very dense control panel. Hand

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Held TVNCRIStereo controls could be ergonomically
formed and allow for more powerful features if this sensor
technology is used.
The sensor of the present invention can be conformed to
any surface and can be made to detect multiple touching
points, making possible a more powerful joystick. The
unique pressure detection ability of the sensor technology of
the present invention is also key to this application. Computer games, "remote" controls (hobby electronics, planes),
and machine tool controls are a few examples of applications which would benefit from the sensor technology of the
present invention.
Musical keyboards (synthesizers, electric pianos) require
velocity sensitive keys which can be provided by the pressure sensing ability of this sensor. There are also pitch
bending controls, and other slide switches that could be
replaced with this technology. An even more unique application comprises a musical instrument that creates notes as
a function of the position and pressure of the hands and
fingers in a very articulate 3-d interface.
The sensor technology of the present invention can best
detect any conducting material pressing against it. By adding a conductive foam material on top of the sensor the
sensor of the present invention may also indirectly detect
pressure from any object being handled, regardless of its
electrical conductivity.
Because of the amount of information available from this
sensor it will serve very well as an input device to virtual
reality machines. It is easy to envision a construction that
allows position-monitoring in three dimensions and some
degree of response (pressure) to actions.
While embodiments and applications of this invention
have been shown and described, it would be apparent to
those skilled in the art that many more modifications than
mentioned above are possible without departing from the
inventive concepts herein. The invention, therefore, is not to
be restricted except in the spirit of the appended claims.
What is claimed is:
1. An object proximity sensor, including:
a touch-sensitive transducer disposed on a substrate, said
touch sensitive transducer including a matrix of row
conductive lines disposed in a first direction and column conductive lines disposed in a second direction
generally perpendicular to said first direction, said row
conductive lines and said column conductive lines
insulated from one another, and an insulating layer
disposed over said row conductive lines and said column conductive lines,- said insulating layer forming a
touch surface, said insulating layer having a thickness
selected to promote capacitive coupling between a
finger placed proximate to the touch surface of said
insulating layer and said row conductive lines and said
column conductive lines;
means for simultaneously injecting electrical charge onto
each of said row conductive lines, and for sensing a
row-sense voltage created on each of said row conductive lines by said electrical charge onto each of said row
conductive lines;
means for simultaneously injecting electrical charge onto
each of said column conductive lines, and for sensing
a column-sense voltage created on each of said column
conductive lines by said electrical charge onto each of
said column conductive lines; and
means for producing a set of object-sensed electrical
signals related to said row-sense voltage and said
column-sense voltage; and

means for processing said set of row electrical signals and
said set of column electrical signals to create a proximity electrical signal proportional to the proximity of
said object to said touch surface.
2. The object proximity sensor of claim 1 wherein said
row conductive lines are disposed on a first face of said
substrate and said column conductive lines are disposed on
a second face of said substrate opposite said first face, said
touch sensitive transducer further including a plurality of
spaced-apart conductive sensor pads disposed in a row and
column matrix pattern on said substrate, each of said sensor
pads connected to a corresponding one of said row conductive lines or column conductive lines.
3. The object proximity sensor of claim 1, further including:
means for sensing a minimum no-object proximate
capacitance from among said row conductive lines, for
sensing a minimum no-object proximate capacitance
from among said column conductive lines, for producing a set of minimum background electrical signals
related thereto; and
means for subtracting said set of minimum background
electrical signals from said set of object-sensed electrical signals.
4. The object proximity sensor of claim 1, further including:
means for producing a set of average no-object-proximate
electrical signals related to an average no-object-proximate capacitance from among said row conductive
lines and an average no-object-proximate capacitance
from among said column conductive lines; and
means for subtracting said set of average no-objectproximate electrical signals from said set of objectsensed electrical signals.
5. The object proximity sensor of claim 1 wherein the
ones of said sensor pads associated with odd numbered ones
of said row conductive lines are disposed along a first set of
column positions and the ones of said sensor pads associated
with even numbered ones of said row conductive lines are
disposed at a second set of column positions offset from said
first set of column positions wherein said sensor pads form
a closely packed repetitive pattern wherein each pad is not
in contact with adjoining pads.
6. A method for providing an electrical signal representative of the position of an object in a two dimensional
sensing plane and of the proximity of the object to the two
dimensional sensing plane, including the steps of:
providing a sensing plane including a matrix of conductors arranged as a plurality of rows and columns of
spaced apart row conductive lines and column conductive lines, said sensing plane having an inherent capacitance on the various ones of said row conductive lines
and column conductive lines, said capacitance varying
with the proximity of an object to said row and column
conductors;
simultaneously generating from among said row conductive lines a first electrical signal proportional to a
no-object-proximate value of said capacitance when no
object is proximate to said sensing plane from among
said row conductive lines;
simultaneously generating from among said row conductive lines a corresponding second electrical signal proportional to the value of said capacitance when an
object is located proximate to but not necessarily in
contact with said sensing plane;
subtracting each of said corresponding first electrical
signals from said second electrical signals to produce a
set of row electrical signals;

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simultaneously generating from among said column condimension comprises separately encoding said set of row
electrical signals into a first digital signal and encoding said
ductive lines a third electrical signal proportional to the
set of column electrical signals into a second digital signal.
no-object-proximate value of said capacitance when no
8. The method of claim 6 wherein the step of encoding
object is proximate to said sensing plane from among
5 said set of row electrical signals and said set of column
said column conductive lines;
electrical signals into electrical signals indicating the posisimultaneously generating for each conductor in the coltion of said object in said row dimension and said column
umn dimensions a corresponding fourth electrical sigdimension comprises separately encoding said set of row
nal proportional to the value of said capacitance when
electrical signals into a first digital signal and encoding said
an object is located proximate to but not necessarily in
10 set of column electrical signals into a second digital signal,
contact with said sensing plane;
and further including the step of encoding said proximity
subtracting each of said corresponding third electrical
electrical signal into a third digital signal.
signals from said fourth electrical signals to produce a
9. The method of claim 6 further including the step of
set of column electrical signals;
providing a signal when said row electrical signal for any of
encoding said set of row electrical signals and said set of 15 said row or column electrical signal exceeds a threshold
column electrical signals into electrical signals indicatvalue.
ing the position of said object in said row dimension
10. The method of claim 6, wherein said first and third
and said column dimension; and
electrical signals are proportional to the minimum no-objectproximate value of said capacitance from among said row
processing said set of row electrical signals and said set of
column electrical signals to create proximity electrical 20 conductive lines and column conductive lines.
11. The method of claim 6, wherein said first and third
signal proportional to proximity of said object to said
electrical signals are proportional to the average no-objectsensing plane.
proximate value of said capacitance from among said row
7. The method of claim 6 wherein the step of encoding
said set of row electrical signals and said set of column
conductive lines and column conductive lines.
electrical signals into electrical signals indicating the position of said object in said row dimension and said column
* * * * *

UNITED STATES PATENT AND TRADEMARK OFFICE

CERTIFICATE OF CORRECTION

Page 1 of 2

PATENT NO.

5.495,077

DATED

February 27, 1996

INVENTOR(S) :

Robelt 1. Miller, Stephen Bisset, Timothy P. Allen, GUnter Steinbach

It is certified that error appears in the above-indentified patent and that said Letters Patent is hereby
corrected as shown below:

Title page, item [*] . "Notice", replace "Dec. 20, 2011" with --August 31, 2013--.

Column 1, line 38, replace "doe" with --does--.
Column 2, line 3, after "are" delete "a".
Column 5, line 65, replace "FIG. 1 a" with --FIG. 1a--.
Column 7, line 22, replace "piez-oelectric" with --piezo-electric--.
Column 7, line 30, replace "a" with --an--.
Column 8, line 24, after "connected" insert --to--.
Column 9, line 10, after "array" insert --10.--.
Column 9, line 22, replace "Integrated" with --integrated--.
Column 9, line 25, replace the first occurrence of "Y" with --X--.
Column 10, line 62, replace "(NO)" with --(AlD)--.
Column 11, line 18, replace "sample hold" with --sample/hold--.
Column 11, line 35, replace "VTHMAX" with

--VTHMAX--'

Column 12, line 26, replace "shoaled" with --shorted--.
Column 13, line 32, replace "a" with --an--.
Column 14, line 10, replace "0 v" with --Ov--.

UNITED STATES PATENT AND TRADEMARK OFFICE

CERTIFICATE OF CORRECTION

Page 2 01'2

PATENT NO.

5,495.077

DATED

February 27. 1996

INVENTOR(S) :

Robert J. Miller. Stephen Bisset. Timothy P. Allen. GUnter Steinbach

It is certified that error appears in the above-indentified patent and that said Letters Patent is hereby
corrected as shown below:

Column 17, line 28, replace "VTHMAX" with --VTHMAX--'
Column 17, line 47, replace "VLBIAS" with --VLBIAS--'
Column 18, line 36, replace "encode" with --encoder--.
Column 19, line 32, replace "200-6" with --220-6--.
Column 19, line 51, replace "(NO)" with --(A/D)--.

Signed and Sealed this
Twenty-second Day of October, 1996
Attest:

BRUCE LEHMAN

Attesting Officer

Commissioner of Patents and Trademarks